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ACPL-782T-000E
Automotive Isolation Amplifier
DataSheet
Lead (Pb) Free RoHS 6 fully compliant
RoHS 6 fully compliant options available; -xxxE denotes a lead-free product
Description
The ACPL-782T isolation amplifier was designed for voltage and current sensing in electronic motor drives and battery system monitoring. In a typical implementation, and motor currents flow through an external resistor and the resulting analog voltage drop is sensed by the ACPL-782T. A differential output voltage is created on the other side of the ACPL-782T optical isolation barrier. This differential output voltage is proportional to the motor current and can be converted to a single-ended signal by using an op-amp as shown in the recommended application circuit. Since common-mode voltage swings of several hundred volts in tens of nanoseconds are common in modern switching inverter motor drives, the ACPL782T was designed to ignore very high common-mode transient slew rates (of at least 10 kV/s). The high CMR capability of the ACPL-782T isolation amplifier provides the precision and stability needed to accurately monitor motor current and DC rail voltage in high noise motor control environments, providing for smoother control (less"torque ripple") in various types of motor control applications. The product can also be used for general analog signal isolation applications requiring high accuracy, stability, and linearity under similarly severe noise conditions. The ACPL-782T utilizes sigma delta (-) analog-to-digital converter technology, chopper stabilized amplifiers, and a fully differential circuit topology. Together, these features deliver unequaled isolationmode noise rejection, as well as excellent offset and gain accuracy and stability over time and temperature. This performance is delivered in a compact, auto-insertable, industry standard 8-pin DIP package that meets worldwide regulatory safety standards. (A gull-wing surface mount option -300E is also available).
Features
* * * * * * * * 2% Gain Tolerance @ 25C 15 kV/s Common-Mode Rejection at VCM = 1000V 30ppm/C Gain Drift vs. Temperature 0.3 mV Input Offset Voltage 100 kz andwidth 0.004% Nonlinearity Compact, Auto-Insertable Standard 8-pin DIP Package 782T Worldwide Safety Approval (pending): - UL 1577 (3750 VRMS/1 min.) and - CSA - IEC/EN/DIN EN 60747-5-5 Qualified to AEC-Q100 Test Guidelines Automotive Operating Temperature -40 to 125C Advanced Sigma-Delta (-) A/D Converter Technology Fully Differential Circuit Topology
782T
* * * *
Applications
* Automotive Motor Inverter Current/Voltage Sensing * Automotive AC/DC and DC/DC converter Current/ Voltage sensing * Automotive attery ECU * Automotive Motor Phase Current Sensing * Isolation Interface for Temperature Sensing * General Purpose Current Sensing and Monitoring
FunctionalDiagram
VDD1 1 VIN+ 2 VIN- 3 GND1 4 IDD1 + SHIELD IDD2 8 VDD2 7 VOUT+ 6 VOUT5 GND2
+ -
The connection of a 0.1 F bypass capacitor between pins 1 and 4, pins 5 and 8 is recommended.
CAUTION: It is advised that normal static precautions be taken in handling and assembly of this component to prevent damage and/or degradation which may be induced by ESD.
PartNumber
ACPL-782T
Option (RoHSCompliant)
-000E -300E -500E
Package
300mil DIP-8
SurfaceMount
X X
Gullwing
X X
Tape&Reel
IEC/EN/DINEN 60747-5-5
X X
X
X
To order, choose a part number from the part number column and combine with the desired option from the option column to form an order entry. Example: ACPL-782T-500E to order product of gullwing SMT DIP-8 package in Tape and Reel packaging with RoS compliant. Option datasheets are available. Contact your Avago sales representative or authorized distributor for information.
PackageOutlineDrawings
ACPL-782T-000E Standard DIP Package
9.80 0.25 (0.386 0.010) 8 7 6 5 DATE CODE EXTENDED DATE CODE 4 1.78 (0.070) MAX. 6.35 0.25 (0.250 0.010) 7.62 0.25 (0.300 0.010)
Dimensions in millimeters and (inches). Note: Floating lead protrusion is 0.5 mm (20 mils) max.
ACPL-782T YYWW EE 1 1.19 (0.047) MAX. 2 3
3.56 0.13 (0.140 0.005)
4.70 (0.185) MAX.
0.51 (0.020) MIN. 2.92 (0.115) MIN. 1.080 0.320 (0.043 0.013) 0.65 (0.025) MAX. 2.54 0.25 (0.100 0.010) 5 TYP. 0.20 (0.008) 0.33 (0.013)
2
GullWingSurfaceMountOption300
LAND PATTERN RECOMMENDATION 9.80 0.25 (0.386 0.010)
8 7 6 5
1.016 (0.040)
ACPL-782T YYWW EE
1 2 3 4
6.350 0.25 (0.250 0.010)
10.9 (0.430)
1.27 (0.050) 1.780 (0.070) MAX. 9.65 0.25 (0.380 0.010) 7.62 0.25 (0.300 0.010)
2.0 (0.080)
1.19 (0.047) MAX.
3.56 0.13 (0.140 0.005) 1.080 0.320 (0.043 0.013) 2.54 (0.100) BSC
Dimensions in millimeters (inches). Tolerances (unless otherwise specified): xx.xx = 0.01 xx.xxx = 0.005 Note: Floating lead protrusion is 0.5 mm (20 mils) max.
0.20 (0.008) 0.33 (0.013)
0.635 0.25 (0.025 0.010) 0.635 0.130 (0.025 0.005)
12 NOM.
LEAD COPLANARITY MAXIMUM: 0.102 (0.004)
Recommended Pb-Free IR Profile
Recommended reflow condition as per JEDEC Standard, J-STD-020 (latest revision). Non-alide Flux should be used.
RegulatoryInformation
The ACPL-782T-000E is pending approval by the following organizations:
UL
Pending approval under UL 1577, component recognition program up to VISO = 3750 VRMS expected prior to product release.
IEC/EN/DIN EN 60747-5-5
IEC 60747-5-5: Pending EN 60747-5-5: Pending DIN EN 60747-5-5: Pending
CSA
Approved under CSA Component AcceptanceNotice #5, File CA 88324.
3
IEC/EN/DINEN60747-5-5InsulationCharacteristics
Description
Installation classification per DIN VDE 0110/1.89, Table 1 * for rated mains voltage 300 Vrms * for rated mains voltage 450 Vrms * for rated mains voltage 600 Vrms Climatic Classification Pollution Degree (DIN VDE 0110/1.89) Maximum Working Insulation Voltage Input to Output Test Voltage, Method b[2] VIORM x 1.875 = VPR, Input to Output Test Voltage, Method a[2] VIORM x 1.6 = VPR, Type and Sample Test, tm = 60 sec, Partial discharge < 5 pC ighest Allowable Overvoltage (Transient Overvoltage tini = 60 sec) Safety-limiting values--maximum values allowed in the event of a failure. * Case Temperature * Input Current[3] * Output Power[3] Insulation Resistance at TS , VIO = 500 V 100% Production Test with tm = 1 sec, Partial discharge < 5 pC VIORM VPR VPR VIOTM TS IS,INPUT PS,OUTPUT RS
Symbol
Characteristic
I-IV I-III I-II 55/125/21 2 891 1670 1426 6000 175 400 600 >109
Unit
VPEAK VPEAK VPEAK VPEAK C mA mW
Notes: 1. Insulation characteristics are guaranteed only within the safety maximum ratings which must be ensured by protective circuits within the application. Surface Mount Classification is Class A in accordance with CECC00802. 2. Refer to the optocoupler section of the Isolation and Control Components Designer's Catalog, under Product Safety Regulations section, (IEC/EN/ DIN EN 60747-5-5) for a detailed description of Method a and Method b partial discharge test profiles. 3. Refer to the following figure for dependence of PS and IS on ambient temperature.
OUTPUT POWER - PS, INPUT CURRENT - IS
800 700 600 500 400 300 200 100 0 0 25 50
PS (mW) IS (mA)
75 100 125 150 TA - CASE TEMPERATURE - C
175
200
4
InsulationandSafetyRelatedSpecifications
Parameter
Minimum External Air Gap (Clearance) Minimum External Tracking (Creepage) Minimum Internal Plastic Gap (Internal Clearance) Tracking Resistance (Comparative Tracking Index) Isolation Group (DIN VDE0109) CTI
Symbol
L(101) L(102)
Value
7.4 8.0 0.5
Units
mm mm mm
Conditions
Measured from input terminals to output terminals, shortest distance through air. Measured from input terminals to output terminals, shortest distance path along body. Through insulation distance conductor to conductor, usually the straight line distance thickness between the emitter and detector. DIN IEC 112/VDE 0303 Part 1
>175
Volts
IIIa
Material Group (DIN VDE 0110)
AbsoluteMaximumRatings
Parameter
Storage Temperature Operating Temperature Supply Voltage Steady-state Input Voltage 2 second Transient Input Voltage Output Voltage Solder Reflow Temperature Profile VOUT
Symbol
TS TA VDD1, VDD2 VIN+, VIN-
Min.
-55 -40 0 -2.0 -6.0 -0.5
Max.
130 125 5.5 VDD1 + 0.5 VDD2 + 0.5
Units
C C Volts Volts Volts Volts
See Package Outline Drawings Section
RecommendedOperatingConditions
Parameter
Ambient Operating Temperature Power Supply Voltage Input Voltage (Accurate & Linear) Input Voltage (Functional)
Symbol
TA VDD1, VDD2 VIN+, VINVIN+, VIN-
Min.
-40 4.5 -200 -2
Max.
125 5.5 200 2
Units
C Volts mV V
Notes
1
5
DCElectricalSpecifications
Unless otherwise noted, all typical and figures are at the nominal operating conditions of VIN+ = 0, VIN- = 0 V, VDD1 = VDD2 = 5 V and TA = 25C; all Min. /Max. Specifications are within the Recommended Operating Conditions. Parameter
Input Offset Voltage
Symbol
VOS
Min.
-2.0 -4.0
Typ.*
0.3
Max.
2.0 4.0
Units
mV mV V/C
TestConditions
TA=25C -40C < TA < +125C, -4.5V < (VDD1, VDD2) < 5.5V
Fig.
1,2
Note
Magnitude of Input Offset Change vs. Temperature Gain Magnitude of VOUT Gain Change vs. Temperature VOUT 200 mV Nonlinearity Magnitude of VOUT 200mV Nonlinearity Change vs. Temperature VOUT 100 mV Nonlinearity Maximum Input Voltage before VOUT Clipping Input Supply Current Output Supply Current Input Current Magnitude of Input ias Current vs. Temperature coefficient Output Low Voltage Output igh Voltage Output Common-Mode Voltage Output Short-Circuit Current Equivalent Input Impedance VOUT Output Resistance Input DC Common-Mode Rejection Ratio
|VOS/TA|
3.0
10.0
3
2
G |G/ G/TA| NL200 |NL200/T|
7.84
8.00 30
8.16
V/V PPM/C
-200 mV < VIN+ < 200 mV, TA = 25C,
4,5,6
3 4
0.0037 0.0002
0.35
% %/C
-200 mV < VIN+ < 200 mV
7,8
5
NL100 |VIN+|MAX
0.0027 308.0
0.2
% mV
-100 mV < VIN+ < 100mV 9
6
IDD1 IDD2 IIN+ |IIN/T| -5
10.86 11.56 -0.5 0.45
16.0 20.0
mA mA A nA/C
VIN+ = 400 mV VIN+ = -400 mV
10 11
7 8 9
VOL VO VOCM |IOSC| RIN ROUT CMRRIN 2.2
1.29 3.80 2.545 18.6 500 15 76 2.8
V V V mA k d
10
11
12
6
ACElectricalSpecifications
Unless otherwise noted, all typicals and figures are at the nominal operating conditions of VIN+ = 0, VIN- = 0 V, VDD1 = VDD2 = 5 V and TA = 25C; all Min./Max. specifications are within the Recommended Operating Conditions. Parameter
VOUT andwidth (-3 d) sine wave. VOUT Noise VIN to VOUT Signal Delay (50 - 10%) VIN to VOUT Signal Delay (50 - 50%) VIN to VOUT Signal Delay (50 - 90%) VOUT Rise/ Fall Time (10 - 90%) Common Mode Transient Immunity Power Supply Rejection
Symbol
W NOUT tPD10 tPD50 tPD90 tR/F CMTI PSR
Min.
50
Typ.*
100 6 2.03 3.47 4.99 2.96
Max.
Units
kz mVRMS
TestConditions
VIN+ = 200mVpk-pk VIN+ = 0.0 V Measured at output of MC34081on Figure 15. VIN+ = 0 mV to 150mV step.
Fig.
12,13
Note
13 4,15
3.3 5.6 9.9 6.6
s s s s kV/s mVRMS
10.0
15.0 170
VCM = 1 kV, TA = 25C With recommended application circuit.
16
14 15
PackageCharacteristics
Parameter
Input-Output Momentary Withstand Voltage Resistance (Input-Output) Capacitance (Input-Output)
Symbol
VISO
Min.
3750
Typ.*
Max.
Units
VRMS
TestConditions
R < 50%, t = 1 min. TA = 25C VI-O = 500 VDC = 1 Mz
Fig.
Note
16,17
RI-O CI-O
>109 1.2
pF
18 18
7
Notes: General Note: Typical values represent the mean value of all characterization units at the nominal operating conditions. Typical drift specifications are determined by calculating the rate of change of the specified parameter versus the drift parameter (at nominal operating conditions) for each characterization unit, and then averaging the individual unit rates. The corresponding drift figures are normalized to the nominal operating conditions and show how much drift occurs as the par-ticular drift parameter is varied from its nominal value, with all other parameters held at their nominal operating values. Note that the typical drift specifications in the tables below may differ from the slopes of the mean curves shown in the corresponding figures. 1. Avago Technologies recommends operation with VIN- = 0 V (tied to GND1). Limiting VIN+ to 100 mV will improve DC nonlinearity and nonlinearity drift. If VIN- is brought above VDD1 - 2 V, an internal test mode may be activated. This test mode is for testing LED coupling and is not intended for customer use. 2. This is the Absolute Value of Input Offset Change vs. Temperature. 3. Gain is defined as the slope of the best-fit line of differential output voltage (VOUT+-VOUT-) vs. differential input voltage (VIN+-VIN-) over the specified input range. 4. This is the Absolute Value of Gain Change vs. Temperature in PPM level. 5. Nonlinearity is defined as half of the peak-to-peak output deviation from the best-fit gain line, expressed as a percentage of the full-scale differential output voltage. 6. NL100 is the nonlinearity specified over an input voltage range of 100 mV. 7. The input supply current decreases as the differential input voltage (VIN+-VIN-) decreases. 8. The maximum specified output supply current occurs when the differential input voltage (VIN+-VIN-) = -200 mV, the maximum recommended operating input voltage. owever, the output supply current will continue to rise for differential input voltages up to approximately -300 mV, beyond which the output supply current remains constant. 9. ecause of the switched-capacitor nature of the input sigma-delta converter, time-averaged values are shown. 10. When the differential input signal exceeds approximately 308 mV, the outputs will limit at the typical values shown. 11. Short circuit current is the amount of output current generated when either output is shorted to VDD2 or ground.
12. CMRR is defined as the ratio of the differential signal gain (signal applied differentially between pins 2 and 3) to the common-mode gain (input pins tied together and the signal applied to both inputs at the same time), expressed in d. 13. Output noise comes from two primary sources: chopper noise and sigma-delta quantization noise. Chopper noise results from chopper stabilization of the output op-amps. It occurs at a specific frequency (typically 400 kz at room temperature), and is not attenuated by the internal output filter. A filter circuit can be easily added to the external post-amplifier to reduce the total RMS output noise. The internal output filter does eliminate most, but not all, of the sigmadelta quantization noise. The magnitude of the output quantization noise is very small at lower frequencies (below 10kz) and increases with increasing frequency. 14. CMTI (Common Mode Transient Immunity or CMR, Common Mode Rejection) is tested by applying an exponentially rising/falling voltage step on pin 4 (GND1) with respect to pin 5 (GND2). The rise time of the test waveform is set to approximately 50 ns. The amplitude of the step is adjusted until the differential output (VOUT+-VOUT-) exhibits more than a 200 mV deviation from the average output voltage for more than 1s. The ACPL-782T will continue to function if more than 10 kV/s common mode slopes are applied, as long as the breakdown voltage limitations are observed. 15. Datasheet value is the differential amplitude of the transient at the output of the ACPL-782T when a 1 Vpk-pk, 1 Mz square wave with 40 ns rise and fall times is applied to both VDD1 and VDD2. 16. In accordance with UL 1577, each optocoupler is proof tested by applying an insulation test voltage 4500 VRMS for 1 second (leakage detection current limit, II-O 5 A). This test is performed before the 100% production test for partial discharge (method b) shown in IEC/ EN/DIN EN 60747-5-5 Insulation Characteristic Table. 17. The Input-Output Momentary Withstand Voltage is a dielectric voltage rating that should not be interpreted as an input-output continuous voltage rating. For the continuous voltage rating refers to the IEC/EN/DIN EN 60747-5-5 insulation characteristics table and your equipment level safety specification. 18. This is a two-terminal measurement: pins 1-4 are shorted together and pins 5-8 are shorted together.
8
VDD1 1 2 0.1 F ACPL-782T 3 4 8 7 6 5
VDD2
+15 V 0.1 F 0.1 F 10 K + 10 K VOUT AD624CD GAIN = 100 0.1 F
0.47 F
0.47 F -15 V
Figure1.InputOffsetVoltageTestCircuit.
0.8 VOS - INPUT OFFSET VOLTAGE - mV VOS - INPUT OFFSET VOLTAGE - mV 0.7 0.6 0.5 0.4 0.3 0.2 -55 -25 5 35 65 TA - TEMPERATURE - C 95 125
0.39 0.38 0.37 0.36 0.35 0.34 0.33 4.5 4.75 5.25 5.0 VDD - SUPPLY VOLTAGE - V 5.5 vs. VDD1 vs. VDD2
Figure2.InputOffsetVoltagevs.Temperature.
Figure3.InputOffsetVoltagevs.Supply.
8.035 8.03 G - GAIN - V/V 8.025 8.02 8.015 8.01 -55 -35 -15 5 25 45 65 TA - TEMPERATURE - C 85 105 125
Figure4.Gainvs.Temperature.
9
VDD1 1 2 ACPL-782T 3 0.01 F 4 5 8 7 6
VDD2
+15 V 0.1 F
+15 V 0.1 F
0.1 F VIN 404 13.2
0.1 F 10 K + 10 K AD624CD GAIN = 4 0.1 F + VOUT AD624CD GAIN = 10 0.1 F
0.47 F
0.47 F -15 V
-15 V 10 K 0.47 F
Figure5.GainandNonlinearityTestCircuit.
8.032
0.03 0.025 NL - NONLINEARITY - %
8.03 G - GAIN - V/V
0.02 0.015 0.01 0.005
8.028 8.026 8.024
vs. VDD1 vs. VDD2 4.5 4.75 5.0 5.25 VDD - SUPPLY VOLTAGE - V 5.5
0 -55
-25
5 65 35 TA - TEMPERATURE - C
95
125
Figure6.Gainvs.Supply.
Figure7.Nonlinearityvs.Temperature.
0.005
4.2
0.004
VO - OUTPUT VOLTAGE - V
NL - NONLINEARITY - %
3.4
2.6 1.8 1.0 -0.5
0.003 vs. VDD1 vs. VDD2 0.002 4.5 4.75 5.0 5.25 VDD - SUPPLY VOLTAGE - V 5.5
VOP VOR -0.3 -0.1 0.1 VIN - INPUT VOLTAGE - V 0.3 0.5
Figure8.Nonlinearityvs.Supply.
Figure9.OutputVoltagevs.InputVoltage.
10
13 IDD - SUPPLY CURRENT - mA
0 -1 -2 -3 -4 -5 -0.6 -0.4 0 -0.2 0.2 VIN - INPUT VOLTAGE - V 0.4 0.6
10
7 IDD1 IDD2 4 -0.5 -0.3 -0.1 0.1 VIN - INPUT VOLTAGE - V 0.3 0.5
Figure10.SupplyCurrentvs.InputVoltage.
IIN - INPUT CURRENT - A
PHASE - DEGREES
Figure11.InputCurrentvs.InputVoltage.
1 0 GAIN - dB -1 -2 -3 -4
50 0 -50
-100
-150 -200 -250
10
100
1000 10000 FREQUENCY (Hz)
100000
-300
10
100
10000 1000 FREQUENCY (Hz)
100000
Figure12.Gainvs.Frequency.
Figure13.Phasevs.Frequency.
5.5 PD - PROPAGATION DELAY - S 4.7 3.9 3.1 2.3 1.5 Tpd 10 Tpd 50 Tpd 90 Trise
-55
-25
35 5 65 TA - TEMPERATURE - C
95
125
Figure14.PropagationDelayvs.Temperature.
11
10 K
VDD1
VDD2
+15 V 0.1 F
0.1 F VIN 0.01 F
1 2 ACPL-782T 3
8 0.1 F 7 2K 6 2K VOUT + MC34081 0.1 F
4
5
10 K -15 V
VIN IMPEDANCE LESS THAN 10 W. Figure15.PropagationDelayTestCircuits.
10 K
78L05 IN OUT 0.1 F 0.1 F 1 2 9V 3 ACPL-782T 6 8 7
VDD2
150 pF +15 V 0.1 F 0.1 F 2K 2K VOUT + MC34081 0.1 F
4
5 150 pF VCM
10 K -15 V
PULSE GEN. + Figure16.CMTITestCircuits.
12
ApplicationInformation PowerSuppliesandBypassing
The recommended supply connections are shown in Figure 17. A floating power supply (which in many applications could be the same supply that is used to drive the high-side power transistor) is regulated to 5 V using a simple zener diode (D1); the value of resistor R4 should be chosen to supply sufficient current from the existing floating supply. The voltage from the current sensing resistor (Rsense) is applied to the input of the ACPL-782T through an RC anti-aliasing filter (R2 and C2). Although the application circuit is relatively simple, a few recommendations should be followed to ensure optimal performance. The power supply for the ACPL -782T is most often obtained from the same supply used to power the power transistor gate drive circuit. If a dedicated supply is required, in many cases it is possible to add an additional winding on an existing transformer. Otherwise, some sort of simple isolated supply can be used, such as a line powered transformer or a high-frequency DC-DC converter. An inexpensive 78L05 three-terminal regulator can also be used to reduce the floating supply voltage to 5 V. To help attenuate high-frequency power supply noise or ripple, a resistor or inductor can be used in series with the input of the regulator to form a low-pass filter with the regulator's input bypass capacitor. As shown in Figure 18, 0.1 F bypass capacitors (C1, C2) should be located as close as possible to the pins of the ACPL-782T. The bypass capacitors are required because of the high-speed digital nature of the signals inside the ACPL-782T. A 0.01F bypass capacitor (C2) is also recommended at the input due to the switched-capacitor nature of the input circuit. The input bypass capacitor also forms part of the anti-aliasing filter, which is recommended to prevent high-frequency noise from aliasing down to lower frequencies and interfering with the input signal. The input filter also performs an important reliability function--it reduces transient spikes from ESD events flowing through the current sensing resistor.
HV+ GATE DRIVE CIRCUIT
+ FLOATING POWER SUPPLY -
***
D1 5.1 V
R2 39
C1 0.1 F
C2 0.01 F
ACPL-782T
MOTOR ***
+ R1 RSENSE
*** HV-
Figure17.RecommendedSupplyandSenseResistorConnections.
13
HV+
POSITIVE FLOATING SUPPLY
C5 150 pF
GATE DRIVE CIRCUIT U1 78L05 IN C1 0.1 F R5 68 OUT
***
R3 10.0 K +5 V +15 V
C8 0.1 F
C2 0.1 F
1
8
C4 0.1 F R1 2.00 K U3 + MC34081 C7 C6 150 pF R4 10.0 K -15 V 0.1 F VOUT
2 C3 0.01 F 3 U2
7
6
R2 2.00 K
MOTOR ***
+
-
4 ACPL-782T *** HV-
5
RSENSE
Figure18.RecommendedApplicationCircuit.
PCBoardLayout
The design of the printed circuit board (PC) should follow good layout practices, such as keeping bypass capacitors close to the supply pins, keeping output signals away from input signals, the use of ground and power planes, etc. In addition, the layout of the PC can also affect the isolation transient immunity (CMTI) of the ACPL-782T, due primarily to stray capacitive coupling between the input and the output circuits. To obtain optimal CMTI performance, the layout of the PC board should minimize any stray coupling by maintaining the maximum possible distance between the input and output sides of the circuit and ensuring that any ground or power plane on the PC board does not pass directly below or extend much wider than the body of the ACPL-782T.
R5 C2 C4
C3
TO VDD1 TO RSENSE+ TO RSENSEFigure19.ExamplePrintedCircuitBoardLayout.
TO VDD2 VOUT+ VOUT-
14
CurrentSensingResistors
The current sensing resistor should have low resistance (to minimize power dissipation), low inductance (to minimize di/dt induced voltage spikes which could adversely affect operation), and reasonable tolerance (to maintain overall circuit accuracy). Choosing a particular value for the resistor is usually a compromise between minimizing power dissipation and maximizing accuracy. Smaller sense resistance decreases power dissipation, while larger sense resistance can improve circuit accuracy by utilizing the full input range of the ACPL -782T. The first step in selecting a sense resistor is determining how much current the resistor will be sensing. The graph in Figure 20 shows the RMS current in each phase of a threephase induction motor as a function of average motor output power (in horsepower, hp) and motor drive supply voltage. The maximum value of the sense resistor is determined by the current being measured and the maximum recommended input voltage of the isolation amplifier. The maximum sense resistance can be calculated by taking the maximum recommended input voltage and dividing by the peak current that the sense resistor should see during normal operation. For example, if a motor will have a maximum RMS current of 10 A and can experience up to 50% overloads during normal operation, then the peak current is 21.1 A (=10 x 1.414 x 1.5). Assuming a maximum input voltage of 200 mV, the maximum value of sense resistance in this case would be about 10 m. The maximum average power dissipation in the sense resistor can also be easily calculated by multiplying the sense resistance times the square of the maximum RMS current, which is about 1 W in the previous example. If the power dissipation in the sense resistor is too high, the resistance can be decreased below the maximum value to decrease power dissipation. The minimum value of the sense resistor is limited by precision and accuracy require40 MOTOR OUTPUT POWER - HORSEPOWER 35 30 25 20 15 10 5 0 0 5 20 10 25 15 MOTOR PHASE CURRENT - A (rms) 30 35 440 V 380 V 220 V 120 V
ments of the design. As the resistance value is reduced, the output voltage across the resistor is also reduced, which means that the offset and noise, which are fixed, become a larger percentage of the signal amplitude. The selected value of the sense resistor will fall somewhere between the minimum and maximum values, depending on the particular requirements of a specific design. When sensing currents large enough to cause significant heating of the sense resistor, the temperature coefficient (tempco) of the resistor can introduce nonlinearity due to the signal dependent temperature rise of the resistor. The effect increases as the resistor-to-ambient thermal resistance increases. This effect can be minimized by reducing the thermal resistance of the current sensing resistor or by using a resistor with a lower tempco. Lowering the thermal resistance can be accomplished by repositioning the current sensing resistor on the PC board, by using larger PC board traces to carry away more heat, or by using a heat sink. For a two-terminal current sensing resistor, as the value of resistance decreases, the resistance of the leads become a significant percentage of the total resistance. This has two primary effects on resistor accuracy. First, the effective resistance of the sense resistor can become dependent on factors such as how long the leads are, how they are bent, how far they are inserted into the board, and how far solder wicks up the leads during assembly (these issues will be discussed in more detail shortly). Second, the leads are typically made from a material, such as copper, which has a much higher tempco than the material from which the resistive element itself is made, resulting in a higher tempco overall. oth of these effects are eliminated when a four-terminal current sensing resistor is used. A four- terminal resistor has two additional terminals that are Kelvin-connected directly across the resistive element itself; these two terminals are used to monitor the voltage across the resistive element while the other two terminals are used to carry the load current. ecause of the Kelvin connection, any voltage drops across the leads carrying the load current should have no impact on the measured voltage. When laying out a PC board for the current sensing resistors, a couple of points should be kept in mind. The Kelvin connections to the resistor should be brought together under the body of the resistor and then run very close to each other to the input of the ACPL-782T; this minimizes the loop area of the connection and reduces the possibility of stray magnetic fields from interfering with the measured signal. If the sense resistor is not located on the same PC board as the ACPL-782T circuit, a tightly twisted pair of wires can accomplish the same thing.
Figure20.MotorOutputHorsepowervs.MotorPhaseCurrentandSupply Voltage.
15
Also, multiple layers of the PC board can be used to increase current carrying capacity. Numerous platedthrough vias should surround each non-Kelvin terminal of the sense resistor to help distribute the current between the layers of the PC board. The PC board should use 2 or 4 oz. copper for the layers, resulting in a current carrying capacity in excess of 20 A.
Note: Please refer to Avago Technologies Application Note 1078 for additional information on using Isolation Amplifiers.
supply current to the gate drive power supply in order to eliminate potential ground loop problems. The only direct connection between the ACPL-782T circuit and the gate drive circuit should be the positive power supply line.
OutputSide
The op-amp used in the external post-amplifier circuit should be of sufficiently high precision so that it does not contribute a significant amount of offset or offset drift relative to the contribution from the isolation amplifier. Generally, op-amps with bipolar input stages exhibit better offset performance than op-amps with JFET or MOSFET input stages. In addition, the op-amp should also have enough bandwidth and slew rate so that it does not adversely affect the response speed of the overall circuit. The postamplifier circuit includes a pair of capacitors (C5 and C6) that form a single-pole low-pass filter; these capacitors allow the bandwidth of the post-amp to be adjusted independently of the gain and are useful for reducing the output noise from the isolation amplifier. Many different op-amps could be used in the circuit, including: TL032A, TL052A, and TLC277 (Texas Instruments), LF412A (National Semiconductor). The gain-setting resistors in the post-amp should have a tolerance of 1% or better to ensure adequate CMRR and adequate gain tolerance for the overall circuit. Resistor networks can be used that have much better ratio tolerances than can be achieved using discrete resistors. A resistor network also reduces the total number of components for the circuit as well as the required board space.
SenseResistorConnections
The recommended method for connecting the ACPL782T to the current sensing resistor is shown in Figure 18. VIN+ (pin 2 of the APCL-782T) is connected to the positive terminal of the sense resistor, while VIN- (pin 3) is shorted to GND1 (pin 4), with the power-supply return path functioning as the sense line to the negative terminal of the current sense resistor. This allows a single pair of wires or PC board traces to connect the ACPL-782T circuit to the sense resistor. y referencing the input circuit to the negative side of the sense resistor, any load current induced noise transients on the resistor are seen as a common-mode signal and will not interfere with the current-sense signal. This is important because the large load currents flowing through the motor drive, along with the parasitic inductances inherent in the wiring of the circuit, can generate both noise spikes and offsets that are relatively large compared to the small voltages that are being measured across the current sensing resistor. If the same power supply is used both for the gate drive circuit and for the current sensing circuit, it is very important that the connection from GND1 of the ACPL782T to the sense resistor be the only return path for
Line 1
Note for the Voltage Divider: V (Line) x [ Rb / (Ra+Rb) ] <= 200 mV + SUPPLY 78L05 IN Ra 0.1 F 39 Rb OUT +5 V 0.1 F 1 ACPL-782T 8 7 6 5 2.00 k 0.1 F 2.0 k 6 5 8
150 pF
+5 V +15 V
10.0 k 0.1 F
2 0.01 F 3 4
7 TL032A 0.1 F VOUT
+ 4
150 pF Line 2 Figure21.Recommendedcircuitforvoltagesensingapplication.
10.0 k -15 V
16
VoltagesensingforDCrailmeasurement
ACPL-782T is a suitable device to measure the DC rail voltage over different potentials. In a DC rail voltage sensing application, the Line1 and Line2 in Figure 21 are the DC lines to be measured.
2 1.5 1 0.5 0 0.5
Dividing error when 1% resistors are used(%)
Dividing ratio error due to the tolerances of the resistors
From a differential calculation, the error in the voltage divider of Ra and Rb is expressed as A/A = Ra/(Ra + Rb) * (Rb/Rb - Ra/Ra) (1) Where A is the ratio of the resistor divider consisting of Ra and Rb. Since the errors of the resistors, Rb/Rb and Ra/Ra are independent to each other, we need to take absolute values in equation (1) to know the maximum possible gain error of the divider and it gives A/A = Ra/(Ra + Rb) * ( |Rb/Rb| + |Ra/Ra|) (2) Figure 22 is the plot of the equation (2) when the resistors have 1% tolerance expressing the relationship between the ratio of Ra to (Ra+Rb) and the possible maximum error of the dividing ratio.
DividerError (%)
0.6
0.7 0.8 Ra/(Ra+Rb)
0.9
1.0
Figure22:DividerError%VsResistorsDivider
150 pF
+ SUPPLY Semitec EC2F103A2-40113 Thermistor IGBT attaching type TH 0.1 F 78L05 +5 V IN OUT ACPL-782T 0.1 F 1 8 2 7 3 6 39 4 1 F R3 5
+5 V 0.1 F 2.0 k 2.0 k 150 pF 6
10.0 k +15 V 8 7 TL032A 0.1 F 10.0 k -15 V VOUT 0.1 F
-
R1 RL = R1//(R2+R3)
R2
+ 54
Note on the thermistor and the RL: Vdd x [RL/(Rth + RL)] x [ R3/(R2 + R3)] <= 200 mV, assuming R2+R3 >> R1 Rth: Resistance of thermistor RL: Linearizing resistor value = R1//(R2+R3)
Figure23.Recommendedcircuitfortemperaturesensingapplication.
17
IsolatedTemperatureSensingusingThermistor
Thermistor is widely used to measure temperatures in most systems application. A galvanic isolation between the potential of the Thermistor and that of the analogto-digital is often required when they are mounted in locations such as high voltage potential, electrically noisy environments, poorly grounded environments, where lack of isolation causes either safety or EMI issues. RL = R1//(R2+R3)=R1(R2+R3)/(R1+R2+R3) R2 and R3 divides the voltage across RL so that the voltage fed into ACPL-782T does not exceed +200 mV. The high impedance characteristic of the input terminals of ACPL782T helps in determining those resistors value since one can select relatively high resistance of R2 and R3 and R1 can be determined easily. If R2+R3 >> R1, RL ~ R1 Dividing ratio ~ R3/(R2+R3) As can be seen from the circuit, one might eliminate R1 and RL~(R2+R3) in this case. An application example with a Thermistor designed for measuring IGT's surface temperatures is shown in Figure 23. Where T is the thermistor and the RL is a resistor for linearization. Suitable RL value is determined from the Thermistor characteristic and the temperature range to measure. Please note that the RL value is the compound value of R1, R2 and R3.
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Avago, Avago Technologies, and the A logo are trademarks of Avago Technologies in the United States and other countries. Data subject to change. Copyright (c) 2005-2009 Avago Technologies. All rights reserved. AV02-1565EN - January 30, 2009

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